Highly integrated television tuner on a single microcircuit

ABSTRACT

A broadband integrated receiver for receiving input signals and outputting composite video and audio signals is disclosed. The receiver employs an up-conversion mixer and a down-conversion mixer in series to produce an intermediate signal. An intermediate filter between the mixers performs coarse channel selection. The down-conversion mixer may be an image rejection mixer to provide additional filtering.

RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.11/376,745, entitled HIGHLY INTEGRATED TELEVISION TUNER ON A SINGLEMICROCIRCUIT, filed Mar. 15, 2006, now U.S. Pat. No. 7,274,410 which isa continuation of U.S. patent application Ser. No. 09/572,393, entitledBROADBAND INTEGRATED TUNER, filed May 16, 2000, now U.S. Pat. No.7,079,195 which is a continuation of U.S. patent application entitledBROADBAND INTEGRATED TUNER assigned Ser. No. 08/904,908, now U.S. Pat.No. 6,177,964, which incorporates by reference and was with U.S. patentapplication Ser. No. 08/426,080 (filed Apr. 21, 1995, entitled HIGHLYINTEGRATED TELEVISION TUNER ON A SINGLE MICROCIRCUIT, now U.S. Pat. No.5,737,035), and this application is a continuation in part of U.S.patent application Ser. No. 08/426,080, now U.S. Pat. No. 5,737,035,(co-pendency with U.S. patent application Ser. No. 08/426,080 beingclaimed through U.S. patent application Ser. No. 11/376,745, U.S. patentapplication Ser. No. 09/572,393, and U.S. patent application Ser. No.08/904,908) and is related to application entitled DUAL MODE TUNER FORCO-EXISTING DIGITAL AND ANALOG TELEVISION SIGNALS assigned Ser. No.08/904,693, now U.S. Pat. No. 6,725,463, and application entitledBROADBAND FREQUENCY SYNTHESIZER assigned Ser. No. 08/904,907, now U.S.Pat. No. 6,163,684, all of which are assigned to a common assignee,which applications are hereby incorporated by reference herein.

TECHNICAL FIELD OF THE INVENTION

This invention relates to television tuner circuits and moreparticularly to a broadband analog television tuner fabricated in amicrocircuit device.

BACKGROUND OF THE INVENTION

One of the most significant costs in television manufacturing is thecost of the tuner. The typical cost of a television (TV) tuner is in theneighborhood of $15.00, which, relative to the cost of the entiretelevision set, is very substantial. Part of the solution to reducingtuner cost is to reduce the number of components in the tuner.

Traditionally, tuners have been comprised of two basic components. Thefirst component performs high frequency to intermediate frequency (RF toIF) conversion. Subsequently, the second component performs IF tobaseband conversion. The TV tuner was originally designed for broadcasttelevision reception within a television set, which is essentially astand-alone unit containing a cathode ray picture tube. So, TV tunerswere originally integral parts embedded in a single-purpose device.

Presently, however, state-of-the-art consumer electronic devices use TVtuners that are not a built-in part of a television set. The tuner is aseparate element that is connected to a cathode ray picture tube at somepoint, but the tuner is not an integral part of the monitor. Forexample, TV tuners may be fabricated on circuit boards and theninstalled in personal computer (PC) systems, thereby allowing the PC tofunction as a television set. These tuners convert a radio frequencytelevision signal into a baseband (or low frequency) video signal whichcan then be passed on to other elements in the PC for video processingapplications.

The circuit component that performs the RF-to-IF conversion typicallycomprises one or two integrated circuits and numerous discreteelements—inductors, capacitors and/or transistors. The IF-to-basebandconversion typically includes another integrated circuit, several filterelements, such as ceramic filters and SAW filters, a series of tuningand control elements, such as resistors and potentiometers, variableinductors and/or capacitors, and some other additional externalcomponents. Thus, the complexity of the tuner is fairly high andtypically there may be between 100 and 200 elements on a circuit board.Furthermore, state-of-the-art TV tuners still require that each tuner bealigned by manual tuning before leaving the factory. This manual tuningis one of the most expensive costs associated with the manufacturingprocess and an important factor in the cost of tuners.

Broadcast television tuners of the past have gone through an evolutionover a period of more than 60 years. The earliest tuners utilized vacuumtube technology and required that the minimum number of vacuum tubespossible be used due to their cost, power consumption and dimensions.Therefore, passive components, such as resistors, capacitors, inductorsand transformers, were used as much as possible in most designs. Thisstyle of design continued until about 1960 when TV tuner components,particularly vacuum tubes, began to be replaced by bipolar and MOStransistors. However, the active device count still defined the cost andsize limits of TV tuners and active device count minimization continued.

In the early 1970's the integrated circuit became viable as an elementin the television tuner and the design techniques were dramaticallychanged. Many functions of the tuner utilizing only one tube ortransistor were being replaced with 4 to 20 individual transistors whichcould perform the same function with better precision, less space, lesspower, less heat generation and lower cost. The introduction of theintegrated circuit was gradual, first encompassing only low frequencyelements and then eventually high frequency active elements.Nonetheless, many passive elements external to the integrated circuitsremained in TV tuner designs.

One advance, the SAW (surface acoustic wave) filter, made a significantchange in that several manually tuned inductors and capacitors could beremoved from the tuners and receive-filtering performance could beimproved within a much smaller space and at reduced cost. However, theSAW filter, which is fabricated on a ceramic substrate, cannot beintegrated on a silicon wafer with the rest of the active circuitry andmust therefore remain a discrete component in the final design. Thetrend of the 1980's was to miniaturize all of the passive components andsimplify their associated manual tuning at the factory. In recent years,TV tuners have been reduced in size from requiring fairly largeenclosures, about 2″×5″×1″, to much smaller enclosures, about ½″×2″×⅜″.There is a high premium placed on small size because TV tuners are beingused in smaller and smaller computers, television sets and VCRs. As theequipment in which tuners are used becomes smaller, the size of the TVtuner must decrease also.

As the size of the tuner goes down, and as tuners are used in a widervariety of devices, cost becomes more critical and must be reduced asmuch as possible in order not to represent a large portion of the finalproduct cost. When a timer is used in a television set, the tuner sizeis less critical because the television set inherently has a large mass.But when a tuner is used in other electronic equipment, space becomes apremium and the footprint of the tuner becomes critical.

Accordingly, it is one object of the invention to provide a TV tunerwhich has a relatively low cost and a small footprint for use on aprinted circuit board.

It is another object of the present invention to provide a TV tuner thatmeets or exceeds the performance of state-of-the-art TV tuners while atthe same time reducing the number of external components needed, therebydecreasing the complexity of the printed circuit board and the amount ofcircuit board area needed by the TV tuner.

It is the further object of the present invention to allow for computercontrol of the TV tuner by a serial bus so that the TV tuner may becontrolled by a microcontroller imbedded in the television set, personalcomputer, or other video device.

It is the further object of the present invention to provide a TV tunerwith computer-controlled output impedance characteristics to accommodatedifferent load specifications.

SUMMARY OF THE INVENTION

These and other problems have been solved by a television tuner thatreceives a broad band of RF signals and converts a desired RF televisionchannel to an IF signal having a picture carrier at 45.75 MHz. Toaccomplish this, an architecture was chosen to perform an up-conversionof the RF input signal to a higher internal frequency, which allows thepresent invention to have minimal filtering on the input stages of thereceiver. The present invention is therefore able to operate withoutvariable-tuned input filtering. This eliminates the need for preciselycontrolled variable tuned filters which must be mechanically alignedduring manufacture and are subject to variation in performance due toage, temperature, humidity, vibration and power supply performance. Thiswas a critical drawback of previous tuners that had to be eliminatedbecause it is a source of tremendous error and distortion, as well ascomplexity.

The present invention allows a wide band of frequencies to enter thefront end of the tuner circuit without removing frequencies in an inputband pass tracking filter. An input filter allows RF signals, typicallyin the range from 55-806 MHz, to enter the circuit while rejecting highfrequency signals above the television band. The input signal thenpasses through a low noise amplifier that controls the input signallevel. Following the input filter and amplifier, the RF signal isconverted to an IF signal in a dual mixer conversion circuit. Theconversion circuit generally up-converts the RF to a first IF signal andthen down-converts the first IF signal to a second IF signal having a45.75 MHz picture carrier.

It is advantageous to have the up-conversion performed on-chip to avoiddrive capability problems associated with high frequency signals andnoise coupling problems resulting from integrated circuit externalinterconnections. Following the up-conversion, a first IF band passfilter performs coarse channel selection. The present invention nextperforms a down-conversion on the output of the first IF filter. Thedown-conversion may be accomplished by an image rejection mixing schemethat provides for a higher level of image rejection than that providedsolely by the first IF filter. The use of an image rejection mixer fordown-converting the first IF signal is optional depending upon thecharacteristics of the first IF filter and its ability to rejectunwanted signals.

The present invention advantageously utilizes much less board space thanprevious designs (on the order of 5% to 10% of the prior art designs)and has the potential to dissipate less power. The present inventionalso advantageously operates on a single voltage level, as opposed totwo or three levels for previous designs.

A further technical advantage of the present invention is that the needfor a metal enclosure is reduced. Integration, by itself, allows forsufficient shielding to meet interference standards. The monolithictelevision (MTV) tuner embodied in the present invention is intended toreplace the TV tuner modules presently used in most broadcast televisionreceiver devices. The level of integration of the present inventiondramatically reduces the cost of the basic TV tuner and enhances itsmanufacturability and reliability. The TV tuner of the present inventionis controlled externally by a computer or controller via a digitalserial bus interface, such as the (I²C) bus defined by PhilipsElectronics N.V. A preferred embodiment of the present inventionprovides an antenna input capable of being connected directly to astandard coaxial cable, thereby allowing both antenna and cabletelevision applications.

A preferred embodiment of the present invention is designed to operateon frequencies used for both over-the-air broadcasts and cabletelevision with National Television Standards Committee (NTSC) encodedvideo. Receiver sensitivity is set to be limited by the antenna noisetemperature for VHF systems. The present invention also employs awide-range automatic gain control (AGC).

For analog television signals, the baseband video output of the presentinvention is leveled, or has minimal variation in video amplitude withrespect to antenna RF signal level. Audio output is broadband compositeto allow connection to an external MTS decoder.

Control is accomplished via a digital service bus interface. The biasand control circuits in a preferred embodiment of the present inventioncontain internal registers which can be updated via the control bus inresponse to changes in operating frequency, transmission standards suchas NTSC, PAL, SECAM and MTS, power, and test modes. Status of the biasand control circuits can be checked via a status register accessiblethrough the I²C bus interface. Status data include AFC error, channellock and received signal strength indicator.

The operating frequency of the present invention is referenced to anexternal crystal or reference frequency generator. A minimum of externalcomponents are used in one embodiment of the present invention to reducethe need for tuning of any components.

The present invention may be implemented in Bipolar, BiCMOS, or CMOSprocesses. However, a preferred embodiment of the present inventionemploys a BiCMOS process to reduce the difficulty in developing thedesign by allowing maximum flexibility.

In the preferred embodiment, the present invention would be constructedentirely on a single integrated substrate. However, design,manufacturing and cost considerations may require that certain elementsbe embodied as discrete off-chip devices.

The foregoing has outlined rather broadly the features and technicaladvantages of the present invention in order that the detaileddescription of the integrated television tuner that follows may bebetter understood. Additional features and advantages of the monolithictelevision tuner will be described hereinafter which form the subject ofthe claims of the invention. It should be appreciated by those skilledin the art that the conception and the specific embodiment disclosed maybe readily utilized as a basis for modifying or designing otherstructures for carrying out the same purposes of the present invention.It should also be realized by those skilled in the art that suchequivalent constructions do not depart from the spirit and scope of theinvention as set forth in the appended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present invention, and theadvantages thereof, reference is now made to the following descriptionstaken in conjunction with the accompanying drawings, in which:

FIG. 1 is a detailed block diagram of the present invention;

FIG. 2 shows the multiple phase lock loop circuit for creating the LOreference signals;

FIG. 3 is a detailed diagram of the phase lock loop circuit; and

FIG. 4 is a detailed block diagram of a state-of-the-art televisiontuner found in the prior art.

DESCRIPTION OF THE PRIOR ART

Before discussing the monolithic television tuner of the presentinvention, it will be useful to discuss a state-of-the-art televisiontuner found in the prior art.

While there have been theoretical proposals to integrate TV tuners in asingle microcircuit, none are known to have been implemented. The nextbest definition of the known prior art, then, is a highly miniaturized,but not fully integrated, tuner as shown in FIG. 4.

FIG. 4 depicts a functional electrical block diagram of a presentstate-of-the-art TV tuner configuration. Television tuner 400 isconstructed in a single metallically shielded assembly containing aprinted circuit board on which all of the associated tuner componentsare mounted and electrically connected. TV tuner 400 is designed to be amodule mounted on other printed circuit boards to allow for directconnection of the input and output signals to their appropriateterminations within the television receiving system. The metal shield isused to keep undesired external signals from interfering with theoperation of the TV tuner 400 and to prevent TV tuner 400 from radiatingsignals that interfere with the operation of external devices.

Prior art television tuner 400 is comprised of three integratedcircuits: preamplifier and mixer 405, IF and baseband signal processor410 and frequency synthesizer and Inter Integrated Circuit (IIC or I²C)bus interface 415. Television tuner 400 is also comprised of a pluralityof discrete components, including bandpass and image reject notch filter404, bandpass and image reject notch filter 412, surface acoustic wave(SAW) filter 416, video carrier filter 424, and audio carrier phaseshifter 460.

Television tuner 400 receives a standard television RF signal fromeither antenna 402 or a cable system connection (not shown) throughbandpass and image reject notch filter 404. Bandpass and image rejectnotch filter 404 limits the signals entering TV tuner 400 so that aminimum number of undesired signals exist in TV tuner 400. Filter 404therefore limits the image response caused by the first mixer, describedlater. Filter 404 also attenuates signals not in a fairly narrow (100MHz) range about the desired signal. Finally, known interferencesignals, such as FM broadcast, shortwave service signals, signals in theintermediate frequency band and Citizen Band radio signals, arespecifically rejected by filter 404.

Preamplifier 406 of preamplifier and mixer 405 receives the output ofbandpass and image reject notch filter 404 and raises the signal level(10 dB) with minimum increase in the noise level (typically 8-10 dB).The gain of preamplifier 406 is controlled by automatic gain control(AGC) 438, so that when a very strong signal enters TV tuner 400,overall gain is reduced, resulting in less distortion in thepreamplifier than without the gain reduction.

The output of preamplifier 406 is sent to bandpass and image rejectnotch filter 412, with the same basic requirement of minimizing thepassage of potential interference signals. Filter 412 is external topreamplifier and mixer 405 and is comprised of a plurality of discreteelements, including capacitors, inductors and varactor diodes.

The output of bandpass and image reject notch filter 412 is then sentback to mixer 408 in preamplifier and mixer 405. Mixer 408 mixes theoutput of filter 412 with the output of a local oscillator, frequencysynthesizer 442, which has a frequency chosen to be higher than thedesired receiver carrier by 45.75 MHz. Thus, the output of mixer 408 is45.75 MHz. There also is an image signal due to mixer 408 at 91.5 MHzabove the input frequency, which is removed by filter 404 and filter412. Therefore, as the frequency of frequency synthesizer 442 is tunedto receive signals of different carrier frequencies, the bandpass andimage reject filters 404 and 412 must also be tuned to properly passonly the desired signals and not the mixer images.

Frequency synthesizer 442 receives an input frequency reference signal(usually 16 bits) and outputs the status signals AUTOMATIC FREQUENCYCONTROL (AFC) ERROR and FREQUENCY (FREQ) LOCK. Additionally, a tuningsignal which is used by the voltage controlled oscillator (VCO) infrequency synthesizer 442 is output from frequency synthesizer 442 tobandpass and image reject notch filters 404 and 412. A local oscillatorsignal is output from frequency synthesizer 442 to mixer 408.

The 45.75 MHz output signal of mixer 408 then passes through SAW(surface acoustic wave) filter 416, which limits the bandwidth of thesignal to only one (1) channel (6 MHz for NTSC standard) and applies alinear attenuation in frequency known as the Nyquist slope around thevisual carrier frequency. The linear attenuation by SAW filter 416converts the signal from a vestigial sideband signal to one which isequivalent to a single sideband with a carrier, so that the frequencyresponse of the signal after demodulation is flat over the videobandwidth. SAW filter 416 is very “lossy” (on the order of 25 dB), sothe input to SAW filter 416 is amplified by a preamplifier (not shown)by a corresponding amount to minimize noise effects.

The output of SAW filter 416 is input to intermediate frequency (IF)amplifier 420 in IF and baseband signal processor 410. IF amplifier 420provides most of the overall gain of TV tuner 400 and receives gaincontrol from AGC 448.

The output of IF amplifier 420 is sent to video detector 422 and is alsosent off-chip to external video carrier filter 424. This is the stage atwhich video demodulation is performed. Video detector 422 is essentiallya mixer with the local oscillator input connected to the output of videocarrier filter 424 through carrier amplitude limiter 426. The output ofthe carrier limiter 426 is an in-phase representation of the videocarrier signal without any modulation applied to it. The output ofcarrier limiter 426 is received by video detector 422, which mixes theoutput of carrier limiter 426 with the output of IF amplifier 420.

AFC frequency discriminator 440 is used in the prior art device todetect the difference between the carrier frequency contained in theoutput of carrier limiter 426 and a known valid carrier frequencyreference. The output signal on the output of AFC frequencydiscriminator 440 is an error signal which is used to drive frequencysynthesizer 442 in a direction that will reduce the error between theoutput of carrier limiter 426 and the known valid carrier frequencyreference. The output of the video detector 422 is a baseband videosignal combined with several high frequency mixing artifacts. Theseartifacts are removed by a video baseband filter 430. The output ofvideo baseband filter 430 is fed to synchronization pulse clamp (syncclamp) 432, which sets the level of the sync pulses to a standard level.

Next, the output of sync clamp 432 is sent to noise invertor 434, whichremoves large noise spikes from the signal. The output of noise invertor434 is sent to video buffer 436, which is configured to drive fairlyhigh circuit board impedances of approximately 1000 to 2000 ohms.

The output of noise invertor 434 is also sent to AGC (automatic gaincontrol) 438, which compares the level of the synchronization pulses tothe signal blanking level to measure the incoming signal strength andgenerates a gain control signal which is used by IF amplifier 420 and RFpreamplifier 406 to dynamically adjust the gain of the TV tuner 400 forthe correct level at the final output.

The audio signal is an FM signal which follows the same path as thevideo through video detector 422. At the output of video detector 422,the audio signal appears as a subcarrier at 4.5 MHz, due to the factthat the audio signal comes into prior art TV tuner 400 4.5 MHz higherin frequency than the desired video carrier. The audio subcarrier ispassed on to an FM quadrature demodulator. The FM quadrature demodulatoris comprised of a mixer, audio second detector 450, and a 90 degree (at4.5 MHz) phase shifter, audio carrier phase shifter 460. The output ofthe audio second detector 450 is a baseband audio signal, which isfiltered by lowpass (30 kHz) filter 452 to remove any undesired highfrequency components. The output of lowpass filter 452 is finally passedon to audio buffer 454, which drives an audio amplifier that ultimatelydrives a speaker. Serial digital interface 444 receives SERIAL DATA andSERIAL CLOCK inputs to provide control and update status for the priorart television receiver.

Baseband and image reject notch filters 404 and 412 are typicallycomprised of a plurality of capacitors, inductors and varactor diodes.Video carrier filter 424 is usually comprised of three discreteelements: an inductor and two capacitors. Likewise, audio carrier phaseshifter 460 is also comprised of an inductor and two capacitors. Inaddition to the circuit elements shown as discrete components outside ofcircuit elements 405, 410 and 415 in FIG. 4, other discrete components(not shown) are connected to IF and baseband signal processor 410 andfrequency synthesizer 442 for tuning purposes. Frequency synthesizer 442is typically tuned by several external capacitors, inductors and/orvaractor diodes. Video buffer 436 and audio buffer 454 will alsotypically employ external discrete elements, such as resistors,capacitors and/or transistors. Video baseband filter 430 and low passfilter 452 may also employ external inductors and capacitors.

DETAILED DESCRIPTION OF THE INVENTION

Turning now to FIG. 1, the preferred embodiment of the present inventionis shown as broadband television tuner 10. The operation of the IFsignal processing components of tuner 10 is further disclosed in theabove-referenced co-pending applications entitled DUAL MODE TUNER FORCO-EXISTING DIGITAL AND ANALOG TELEVISION SIGNALS, and INTERFERENCE-FREEBROADBAND TELEVISION TUNER and BROADBAND FREQUENCY SYNTHESIZER. RFsignals are received in tuner 10 through input filter 101 which has ahigh dynamic range and good linearity across the television frequencyband. Filter 101 operates to attenuate signals above an input cutofffrequency corresponding to the highest frequency in the television band.As distinguished from the prior art, filter 10 is not a narrow band passtracking filter which attenuates most television channels from thereceived signal. Instead, filter 101 passes all channels in thetelevision band.

Following filter 101, the RF signal passes through delayed AGC amplifier102 which operates in conjunction with IF AGC amplifier 116 to controlthe overall signal level in tuner 10. Amplifier 102 may be a variablegain amplifier or a variable gain attenuator in series with a fixed gainamplifier. The preferred embodiment of amplifier 102 comprises a lownoise amplifier (LNA) with a high linearity that is sufficient to passthe entire television band. Amplifier 102 functions to control highinput signal levels in the received RF signal. Tuner 10 is capable ofreceiving signals from a variety of sources, such as an antenna or acable television line. The cable television signals may have a signalstrength of +15 dBmV and may comprise 100 cable channels. Amplifier 102regulates the varying signal levels in this broad band of receivedchannels.

Mixer 103 receives inputs from amplifier 102 and local oscillator 104. Afirst IF signal is generated in mixer 103 and provided to first IFfilter 109. Filter 109 is a band pass filter that provides coarsechannel selection in tuner 10. As a matter of design choice, filter 109may be constructed on the same integrated circuit substrate as mixers103 and 110 or filter 109 may be a discrete off-chip device. Filter 109selects a narrow band of channels or even a single channel from thetelevision signals in the first IF signal.

Following IF filter 109, mixer 110 mixes the first IF signal with asecond local oscillator signal from local oscillator 111 to generate asecond IF signal. Mixer 110 may be an image rejection mixer, ifnecessary, to reject unwanted image signals. The characteristics offirst IF filter 109 will determine whether mixer 110 must provide imagerejection. If the image frequencies of the desired channel areadequately attenuated by first IF filter 109, then mixer 110 may be astandard mixer.

Local oscillators 104 and 111 are controlled by tuning phase locked loopcircuit 105. In the preferred embodiment, the local oscillatorfrequencies are selected so that the picture carrier of a particularchannel in the RF signal will appear at 45.75 MHz in the second IFsignal. However, it will be understood that the present invention is notlimited to specific IF or LO frequencies. Tuning PLL circuit 105receives reference signals from reference oscillator 106 which is drivenby 5.25 MHz crystal 107. I²C 108 provides control inputs and monitorsthe status of tuner 10 and tuning PLL circuit 105.

In operation, the front end of tuner 10 receives the entire televisionband through filter 101 and amplifier 102. Following mixer 103, the RFinput is converted so that a selected channel in the RF signal appearsat a first IF frequency that is selected to pass through filter 109. Thefirst IF frequency is then converted to a second IF frequency of 45.75MHz at the output of mixer 110. The frequencies of the first and secondlocal oscillator signals will vary depending upon the specific channelin the RF signal that is desired. In the preferred embodiment, the firstlocal oscillator frequency is selected so that mixer 103 performs anup-conversion of the RF signal. Following filter 109, the first IFsignal is then down-converted to 45.75 MHz in mixer 110.

Following mixer 110, the second IF signal is further processed by eitherdigital or analog circuits. Second IF filter 113 may be constructed onthe same integrated circuit substrate as the other elements of tunercircuit 10 or it may be a discrete off-chip device. When second IFfilter 113 is a discrete off-chip element, then amplifiers 112 and 114are used to provide proper impedances for filter 113 as well as toprovide gain to maintain system noise performance. After amplifier 114,the signal either remains on-chip for further processing or it can beprovided to an off-chip device, such as a decoder (not shown), throughbuffer 115.

If the signal is processed on-chip, then the second IF signal passesthrough IF AGC amplifier 116 which operates in conjunction with delayedAGC amplifier 102 to control the overall tuner gain. One output ofamplifier 116 is provided to coherent oscillator (COHO) circuit 118.COHO 118 generates two reference signals, one that is in-phase with the45.75 MHz second IF signal and another that is 90° out-of-phase with thesecond IF signal. A third output from COHO 118 is provided to frequencydiscriminator 120 which monitors the frequency of the signal that isprocessed in COHO 118 and generates a tuning error signal for I²Ccontrol 108.

AGC amplifier 116 also drives video detector 121 and audio detector 122.Video detector 121 mixes the second IF signal with the in-phasereference signal from COHO 118. AGC circuit 127 monitors the output ofvideo detector 121 and adjusts the gain of amplifiers 102 and 116 inorder to control the overall tuner gain. If an off-chip decoder isconnected to tuner 10 through buffer 115, then the decoder can controlthe signal gain by providing an input directly to AGC 127.

The signal from video detector 121 passes through sound trap 124 whichremoves the audio carrier from the signal. The output of sound trap 124drives noise clipping circuit 125 which removes large noise spikes whichmay be present in the video signal. Finally, a composite video signal isprovided through buffer 126.

Audio detector 122 mixes the second IF signal with the 90° out-of-phaseor quadrature signal from COHO 118. The output signal from audiodetector 122 will contain an audio carrier at 4.5 MHz and a chromacarrier at approximately 3.6 MHz. Chroma reject filter 129 is a highpass filter that removes the picture and chroma carriers from the outputof audio detector 122. The remaining audio signal is then mixed with a5.25 MHz reference signal in mixer 130 to create a 750 KHz output. Soundfilter 131 is a band pass filter that further filters the 750 KHz audiosignal. FM demodulator 132 is a delay line type of demodulator whichcreates a standard composite audio signal from the 750 KHz FM audiosignal. This audio signal is then provided as the composite audio signalthrough buffer 133.

In an alternative embodiment of the present invention, a plurality oftuners 10 are placed on a single integrated substrate and a single RFinput drives the plurality of tuners 10. This allows a single integrateddevice to concurrently provide different television channels through theoutput of each tuner. This embodiment could be used to drive a“picture-in-a-picture” display or any other display format that requiresmultiple tuners. In another alternative embodiment, the plurality oftuners on a single substrate are coupled to independent RF signalsources and provide independent television signals.

The present invention can be used in applications other than aconventional television receiver. Tuner 10 can be embodied as part of an“add-in” board or a component of a personal computer. This allows a userto receive and view television signals on the computer's display. Theuser could also record or capture television programs directly to thecomputer's memory. The computer could then be used to replay recordedprograms or to manipulate or alter selected frames or segments of thecaptured video and audio signal, or the computer may capture data whichmay have been imbedded in the video signal.

Furthermore, the present invention will be understood to not be limitedto an integrated substrate. Prior art tuners require the use of anarrow-band, tunable filter to eliminate undesired channels from thereceiver. The present invention is distinguished over the prior art byallowing all frequencies in a desired band to enter the front-end oftuner 10 and by removing undesired channels through filtering of the IFsignal.

FIG. 2 shows multiple phase locked loop (PLL) circuits which are used todrive voltage controlled oscillators (VCOs) in order to generate the LOsignals for a dual mixer conversion circuit.

Conversion circuit 20 has dual mixers 202 and 204 which receive LOsignals LO1 and LO2 on lines A and B from local oscillator circuit 20.

In a television system, signals representing individual channels areassigned to specific frequencies in a defined frequency band. Forexample, in the United States, television signals are generallytransmitted in a band from 55 MHz to 806 MHz. The received RF signalspass through a front-end filter 200. In the prior art, filter 200usually was a bandpass tracking filter that allowed only a narrow rangeof frequencies to pass. In the preferred embodiment, filter 200 is a lowpass filter that is designed to remove all frequencies above an inputcutoff frequency. The input cutoff frequency is chosen to be higher thanthe frequencies of the channels in the television band. The output offilter 200 then passes through amplifier 201 to adjust the signal levelthat is provided to mixer 202. When conversion circuit 20 is used in areceiver circuit, amplifier 201 may be an automatic gain control (AGC)amplifier that is adjusted to maintain an overall receiver gain.Following amplifier 201, the RF signal is provided to mixer 202 where itis mixed with a local oscillator signal LO1 from local oscillatorcircuit 30. The output of mixer 202 is first intermediate frequencysignal IF1. Typically, the frequency of LO1 is variable and will beselected based upon the channel in the RF signal that is being tuned.LO1 is selected so that mixing of LO1 and RF in mixer 202 generates anIF1 signal either at a specified frequency or within a narrow range offrequencies.

Following mixer 202, IF filter 203 is a band pass filter that is used toremove unwanted frequencies and spurious signals from the IF1 signal.The band of frequencies that are passed by filter 203 is a matter ofdesign choice depending upon the IF1 frequency selected in eachparticular conversion circuit. In the preferred embodiment, IF filter203 is centered at 1090 MHz and has a 14 MHz pass band. This allows theselected IF1 frequency to vary within 1083-1097 MHz. Mixer 204 receivesboth the filtered IF1 signal from filter 203 and a second localoscillator signal (LO2) from oscillator circuit 20. These signals aremixed to generate a second intermediate frequency (IF2) at the output ofmixer 204. In the preferred embodiment, mixer 204 is an image rejectionmixer that rejects image frequencies from the IF2 signal. LO2 may be avariable or fixed frequency depending upon whether IF1 is at a fixedfrequency or if it varies over a range of frequencies. In either case,the frequency of LO2 is selected to generate a fixed frequency IF2signal. The IF2 signal is provided through amplifiers buffer 205 toadditional processing circuitry (not shown) to generate either digitalor analog television signals. In the preferred embodiment, the frequencyof IF2 is selected to be 45.75 MHz.

An additional consideration when using a dual mixer conversion circuitin a television receiver is the relationship of the picture, chroma andaudio carriers in an analog television signal. This is discussed in theabove-referenced applications.

For analog television signals, it is desirable to choose a combinationof LO1 and LO2 so that the relationship between the picture, chroma andaudio carriers is always the same in the IF2 signal. When the IF2 signalis further processed after amplifier 205, it may be a consideration thatthe analog processing circuits are able to find the chroma and audiocarriers in the same place, either above or below the picture carrier,for every channel. In the preferred embodiment, LO1 and LO2 are selectedso that the IF2 spectral relationship is the inverse of the RF spectralrelationship. That is, the picture carrier is converted from an RFsignal of 55-806 MHz to an IF2 signal at 45.75 MHz with the audiocarrier 4.5 MHz below the picture carrier and the chroma carrier 3.6 MHzbelow the picture carrier.

The audio and chroma carriers are below the picture carrier frequency.This is accomplished by using the lower LO2 frequency (1041 MHz) withthe higher LO1 frequency (1160.25 MHz) or using the higher LO2 frequency(1137.5 MHz) with the lower LO1 frequency (1018.5 MHz).

LO1 is generated in local oscillator circuit 30 (FIG. 3) by PLL1 31 andLO2 is generated by PLL2 32. PLL3 33 and PLL4 34 provide referenceinputs to PLL2 32. I²C 320 controls local oscillator circuit 30 andcauses PLL1-4 31-34 to select the correct LO1 and LO2 frequencies. Localoscillator circuit 30 receives reference signals from oscillator 222 andreference frequency generator 223. Oscillator 222 provides a 5.25 MHzoutput based on crystal 221. Frequency generator 223 divides the 5.25MHz signal from oscillator 222 to generate additional reference signalsat other frequencies.

Local oscillator circuit 30 and PLL1-4 31-34 are shown in greater detailin FIG. 3. PLL1 31 provides the first local oscillator signal (LO1) tomixer 302. PLL2 32, PLL3 33 and PLL4 34 cooperate to provide the secondlocal oscillator signal (LO2) to mixer 204. PLL1 31 receives a 5.25 MHzreference signal at phase comparator 305. The output of phase comparator905 feeds loop amplifier 302 which, in turn, provides the input for VCO1901. There are two outputs from VCO1 301. One output provides the LO1signal to mixer 202 over line A. The other output goes into a dividernetwork comprised of ÷8/÷9 circuit 303 and ÷N circuit 304. Dividercircuits 303 and 304 divide the output of VCO1 301 down to a signalhaving a frequency of 5.25 MHz. This divided-down signal is comparedwith the 5.25 MHz reference signal in phase comparator 305 to completethe phase locked loop.

The output of VCO1 31 is variable between 1145-1900 MHz on the high sideand 572-1033 MHz on the low side. Frequencies below 572 MHz are not usedin LO1 to minimize the introduction of interference frequencies into theconversion circuit. LO1 is chosen from within these ranges so that IF1signal is within the 1090 MHz±7 MHz pass band of filter 303. The 5.25MHz reference signal creates an output stepsize of 5.25 MHz in LO1 whichis utilized for coarse tuning in conversion circuit 10. In the preferredembodiment, PLL1 31 has a bandwidth on the order of 500 KHz. A widebandwidth is preferable to get good close-in phase noisecharacteristics.

Fine tuning (for example to the exact desired channel) is accomplishedby LO2 which is produced by the operation of 3 phase lock loops PLL2 32,PLL3 33 and PLL4 34. PLL4 34 has the same basic configuration as PLL131. It has reference signal of 2.625 MHz which is input to phasecomparator 335. The output of phase comparator 335 drives loop amplifier332 which in turn drives VCO4 331. The output of VCO4 331 has frequencyrange of 220-440 MHz with a 2.625 MHz stepsize and is provided to twodivider circuits. One output of VCO4 331 goes to a divider networkcomprised of ÷6/÷7 circuit 933 and ÷N circuit 334. The effect of dividernetwork 333 and 334 is to divide the output signal of VCO4 330 back downto 2.625 MHz. This signal is then compared with the 2.625 MHz referencesignal in phase comparator 335 to complete the phase locked loop. Theother output of VCO4 331 is provided to ÷42 circuit 330. The output ofdivider 330 is a signal with a frequency range of 5.25-10.5 MHz andhaving a 62.5 KHz stepsize. The output of divider 330 serves as areference signal for PLL2 32.

In PLL3 33, a 5.25 MHz reference signal is input to phase detector 322.Phase detector 322 drives loop amplifier 321 which in turn drives VCO3320. The output of VCO3 33 is divided back down to 5.25 MHz by ÷Ncircuit 323 and then fed back into phase detector 322 to complete theloop. The output of VCO3 33 is selectable between 1128.75 MHz and1034.25 MHz. The selection between these two frequencies determineswhether LO2 is on the high side or the low side.

In PLL2 32, the signal from PLL4 34 is received by phase comparator 314which in turn drives loop amplifier 313. The output of loop amplifier313 controls VCO2 310. VCO2 310 provides the LO2 signal for mixer 204over line B. The LO2 signal varies between 1134.75-1140.125 MHz on thehigh side and 1039.875-1045.125 MHz on the low side. Another output fromVCO2 310 passes through buffer amplifier 311 and then drives imagereject mixer 312. Mixer 312 receives its other input from PLL3 33. Sincethe signal from PPL3 33 is near the frequency of the LO2 signal in VCO2310, it is important that the reverse isolation between mixer 312 andVCO2 310 is good to prevent the PLL3 33 signal from passing into the LO2output of VCO2 310. The output of mixer 312 is provided to phasecomparator 314 to complete the loop in PLL2 32.

In the preferred embodiment, the loop bandwidths of PLL2 32, PLL3 33 andPLL4 34 are all wide to provide good overall close-in phase noise. PLL232 and PLL3 33 have bandwidths of approximately 300-500 KHz. Thebandwidth of PLL4 34 is approximately 200-300 KHz. These bandwidths givephase noise at 100 KHz that is satisfactory for most applications.

The architecture of the frequency synthesis system provides for severalbenefits with respect to the overall operation of the tuner system.These benefits are in providing a lower distortion detection means,immunity to injection locking, a frequency synthesis system that allowsfor wide bandwidth PLLs while preserving a small step size, andproviding for a choice of reference frequency that is out-of-band andthat can be directly used to down-convert the audio portion of thedesired channel.

A wide loop bandwidth for LO1 and LO2 is preferred because this yieldsgood close-in phase noise characteristics for these two signals. This isimportant because it allows the COHO to have a narrow loop bandwidth,which yields a lower distortion video detector. For example, certaincontent within the video signal, such as the horizontal sync signal atapproximately 15 KHz, would be partially tracked by a wide band COHOleading to distortion in the detection process. If the bandwidth of theCOHO is less than 15 KHz, then the COHO would not partially track thehorizontal sync signal leading to a near distortion free detectionprocess. In the prior art, the oscillators used for conversion to IFtypically do not have good close-in phase noise characteristics,requiring a COHO with wide loop bandwidth to track out this noise. It isthus typical in the prior art to employ wider bandwidth COHO's, whichhave the undesirable trait of partially tracking strong signals in thevideo signal, such as horizontal sync, leading to distortion in thedetection process.

It is generally known that the immunity of a phase locked loop toinjection locking is determined by the product of the quality factor, Q,of the VCO and the loop bandwidth. For the case of a VCO implemented ona single chip, it is typically difficult to realize high Qs. Thisconflicts with the integrated circuit implementation of a RF system withPLLs in that the other circuitry sharing the common substrate is asource of spurs that then may be passed on to the PLLs output or lead toinjection locking by the PLL. Since a high Q VCO is not feasible withoutexternal components, the benefit of a wide loop bandwidth of the PLL ismore pronounced.

It is typical in the prior art to make the PLL reference frequency equalto the step size of the frequency synthesizer system. It is furthertypical of the prior art to employ a single loop frequency synthesizerto create the first LO in tuners. For example, if the step size of thesystem was 62.5 KHz, then the reference frequency to the single loop PLLwould also be 62.5 KHz. It is highly desirable to suppress harmonics andspurs of the reference that are in band to a level below the noise floorof the VCO, requiring the loop bandwidth of the PLL to be less than thereference frequency. In the case where the reference is the step size,the loop bandwidth is rather narrow. Consequently, is a clear advantageof the frequency synthesizer described herein to provide both a smallstep size as well as a wide bandwidth for LO1 and LO2 providing forenhanced immunity to spurs as well as providing for a narrow bandwidthCOHO.

A further advantage of the frequency synthesis system is that it can usea reference that is above the baseband frequencies. An example of such afrequency is 5.25 MHz. It should be noted that this 5.25 MHz referenceis above the baseband signal of the system, thus avoiding in-band noiseproduced by the reference and its harmonics. A further advantage of thischoice of reference is that it can be used directly by the audiosubsystem to down convert the frequency modulated audio signal to alower frequency usable by the sound filter and FM demodulator in theaudio subsystem. This eliminates the need for a PLL to create thisfrequency.

Although the present invention and its advantages have been described indetail, it should be understood that various changes, substitutions andalterations can be made herein without departing from the spirit andscope of the invention as defined by the appended claims. Moreover, thescope of the present application is not intended to be limited to theparticular embodiments of the process, machine, manufacture, compositionof matter, means, methods and steps described in the specification. Asone of ordinary skill in the art will readily appreciate from thedisclosure of the present invention, processes, machines, manufacture,compositions of matter, means, methods, or steps, presently existing orlater to be developed that perform substantially the same function orachieve substantially the same result as the corresponding embodimentsdescribed herein may be utilized according to the present invention.Accordingly, the appended claims are intended to include within theirscope such processes, machines, manufacture, compositions of matter,means, methods, or steps.

1. A television receiver comprising: a receiver input coupled to an RFsignal source; a first reference signal having a first operatingfrequency; a first mixer having a first input coupled to said receiverinput and a second input coupled to said first reference signal; abond-wire parallel Inductive Capacitive (LC) filter coupled to an outputof said first mixer; the bond-wire parallel LC filter passing more thanone channel and performing partial image rejection as well as limitingoverall signal power levels to be processed by subsequent circuitry,wherein the more than one channel passed by the bond-wire parallel LCfilter is attenuated; a first and a second phase locked loop providingreference inputs for a third phase locked loop; a second referencesignal produced by said third phase locked loop having a secondoperating frequency; and an image rejection mixer having a first inputcoupled to an output of said bond-wire parallel LC filter and a secondinput coupled to said third phase locked loop to receive said secondreference signal.
 2. The television receiver of claim 1 wherein saidfirst mixer, said bond-wire parallel LC filter, and said image rejectionmixer comprise a differential mixer chain.
 3. The television receiver ofclaim 1 wherein said first mixer and said image rejection mixer areconstructed on a same integrated circuit substrate.
 4. The televisionreceiver of claim 1 wherein said bond-wire parallel LC filter isexternal to an integrated circuit on which said first mixer is located.5. The television receiver of claim 1 wherein said first bandpass filteris external to an integrated circuit on which said first mixer islocated.
 6. A television receiver comprising: a receiver input forreceiving an RF signal; an input filter coupled to said receiver inputand operating to remove all frequency components in said RF signal abovean input cutoff frequency to produce a filtered RF signal; a firstreference signal having a first operating frequency; a first mixerhaving a first input coupled to an output of said input filter and asecond input coupled to said first reference signal; a digital interfaceproviding control with respect to said first mixer; wherein said digitalinterface comprises an inter-integrated circuit interface; a secondreference signal having a second operating frequency; and a second mixerhaving a first input coupled to an output of said first mixer and asecond input coupled to said second reference signal; and a filterbetween said first and second mixers; wherein said second mixer is animage rejection mixer, wherein said first mixer and said second mixerare physically located in the same integrated circuit substrate.
 7. Thetelevision receiver of claim 6 wherein said first mixer, said filter andsaid second mixer comprise a differential mixer chain.
 8. The televisionreceiver of claim 6 wherein said first and second mixers are constructedon a same integrated circuit substrate.
 9. The television receiver ofclaim 6 wherein said filter comprises: a bond-wire parallel InductiveCapacitive (LC) filter.
 10. A method of processing a received RF signal,the method comprising the steps of: attenuating frequency components ofsaid RF signal above a cut-off frequency to create a filtered RF signal;mixing the RF signal with a first reference signal having a firstoperating frequency to thereby produce a first IF signal; controlling afrequency of said first operating frequency for said mixing the filteredRF signal using a digital interface; filtering the first IF signal in abandpass filter to produce an output passband signal having a pluralityof channels; and mixing the output passband signal with a secondreference signal having a second operating frequency to thereby producea second IF signal, wherein said mixing the filtered RF signal and saidmixing the output passband signal are performed in a single integratedcircuit.